1. Field of the Invention
The present invention relates to active power factor correction (PFC). More particularly, the present invention relates to critical mode (CRM) active power factor correction.
2. Description of the Related Art
Most active PFC circuits are based on a boost converter topology. Its main power switch is typically a high-voltage power metal oxide semiconductor field effect transistor (MOSFET) operating at high switching frequency (normally higher than the audible frequency limit of 20 kHz) such that the circuit size can be reduced. There are three major types of active PFC circuits. The first type is continuous current mode (CCM) PFC. The second type is discontinuous current mode (DCM) PFC. The third type is critical mode (CRM) PFC.
FIG. 1A is a schematic diagram showing a conventional CCM PFC circuit 100. Circuit 100 comprises a bridge rectifier DR1-DR4 to rectify the alternating current (AC) line input voltage Vin. The rest of circuit 100 is a boost converter. Circuit 100 operates at a constant switching frequency. The control circuit of power MOSFET Q1 modulates the AC line input current (which is equal to the current through the input inductor L1) such that its waveform follows the rectified sinusoidal AC line input voltage waveform, thereby correcting the power factor.
FIG. 2A is a plot of the current through input inductor L1 versus time. The input inductor current 211 is continuous. It only drops to zero at phase angles of 0° and 180° of the rectified line cycle (typically at 100 Hz or 120 Hz). During each switching cycle of MOSFET Q1, for example, the cycle from T1 to T3, MOSFET Q1 is turned on in the shaded area and is turned off in the non-shaded area. When Q1 is turned on, inductor current 211 rises up. When Q1 is turned off, inductor current 211 falls down. Circuit 100 arranges the relative lengths of the turn-on and turn-off periods of MOSFET Q1 such that the zig-zag waveform of inductor current 211 approximates a sinusoidal waveform which follows the rectified AC input voltage Vin. As shown in FIG. 2A, the average inductor current of circuit 100 is Ipk*sin(α), wherein Ipk is the peak inductor current and α is the phase angle of the rectified line cycle.
For modulating the inductor current, a flip-flop 106 drives MOSFET Q1. Clock signal CLK sets the output of flip-flop 106 to turn on MOSFET Q1 at the beginning of each switching cycle. On the other hand, pulse width modulator PWM resets the output of flip-flop 106 to turn off MOSFET Q1 in each switching cycle. Pulse width modulator PWM compares the output of current sense amplifier 101 and multiplier 104. Multiplier 104 receives three input signals. The first input signal a is a fraction of the rectified AC line input voltage. The second input signal b is the square of input signal a provided by V-square calculator 103. The third input signal c comes from feedback error amplifier 107 and is the amplified difference between reference voltage VREF and a fraction of output voltage Vout. Multiplier 104 outputs a sinewave reference signal Vsin. As shown in FIG. 1A, Vsin=(a*c/b). Sinewave reference signal Vsin represents the upper limit of the inductor current in each switching cycle of MOSFET Q1. When the output of current sense amplifier 101 exceeds sinewave reference signal Vsin, pulse width modulator PWM resets the output of flip-flop 106 to turn off power MOSFET Q1.
The conduction loss of power MOSFET Q1 is minimized since a CCM PFC circuit has the lowest root-mean-square (RMS) current value among all active PFC circuits based on a boost converter. Nevertheless, a CCM PFC circuit suffers from high switching loss due to the slow reverse recovery of the output rectifier diode. In general, the switching loss associated with the recovery time of the output diode is 2% or more of the rated output power.
Further, a CCM PFC circuit has significant switching loss associated with the output capacitance of the power MOSFET. Since the inductor current is continuous, the MOSFET turns on when the output diode is conducting current, and the drain-to-source voltage is at 400V level. The switching loss associated with the MOSFET output capacitance can be as much as 2% of the rated output power. In addition, the reverse recovery of the output diode and the turn-on of the power MOSFET at a high drain-to-source voltage level often lead to severe electromagnetic interference (EMI) issues.
The second major type of PFC is DCM PFC. FIG. 2B is a plot of the inductor current 221 of a typical DCM PFC circuit versus time. DCM PFC operates at constant switching frequency. DCM PFC has no diode reverse recovery issue because the power MOSFET switch always turns on when inductor current 221 is zero. The average inductor current in FIG. 2B is less than 0.5*Ipk*sin(α) due to the dead time from the time the inductor current drops to zero (at T3, T6, T9, etc.) to the time the power switch turns on again (at T4, T7, T10, etc.).
DCM PFC has higher conduction loss than the CCM PFC due to its triangular inductor current waveform. Another issue of DCM PFC is higher order (third order and above) harmonic distortion at light load, especially at high line conditions. The dead time of inductor current 221 accounts for the input current waveform distortion and high-order harmonics.
The third major type of PFC is CRM PFC. FIG. 1B is a schematic diagram showing a conventional CRM PFC circuit 150. FIG. 2C is a plot of the inductor current 231 of CRM PFC circuit 150 versus time. Similar to a DCM PFC circuit, a CRM PFC circuit is a zero-current switching boost converter. The power MOSFET Q1 turns on at zero diode current. Therefore, it eliminates the diode reverse recovery issue. The difference between CRM and DCM is that power MOSFET Q1 of CRM PFC circuit 150 turns on as soon as the current through output diode D1 drops to zero, as shown in FIG. 2C. Whereas, in a DCM PFC circuit, there is a delay time from the time when the output diode current drops to zero to the time the MOSFET turns on. The average inductor current of circuit 150 is 0.5*Ipk*sin(α).
The controlling of power MOSFET Q1 in circuit 150 is similar to that in circuit 100. When the current through output diode D1 drops to zero, the output of zero current detector 111 sets the output of flip-flop 106 to turn on power MOSFET Q1. Sinewave reference signal Vsr provided by multiplier 112 represents the upper limit of inductor current 231. The signal from sensor resistor RSEN to pulse width modulator PWM is proportional to inductor current 231. When the signal from RSEN exceeds sinewave reference signal Vsr, pulse width modulator PWM resets the output of flip-flop 106 to turn off power MOSFET Q1.
CRM PFC does not operate at a constant frequency. Its switching frequency is always higher at both sides of the line cycle (near 0° and 180°) than in the middle of the line cycle (near 90°). At no load or light load conditions, the switching quickens because of small inductor current, therefore increasing switching loss. Such switching loss may overwhelm the output power at light load conditions due to very high switching frequency. In another word, the power efficiency of a CRM PFC circuit is significantly lower at light load conditions. At the present time power supply units of personal computers may be required to conform to the green mode standard. That is, the power consumption of a power supply unit at no load conditions must be lower than a certain threshold, for example, 0.5 W. A high switching loss is undesirable.